PHY Layer Parameter for Body Area Network (BAN) Devices

ABSTRACT

In at least some embodiments, a communication device includes a transceiver with a physical (PHY) layer. The PHY layer is configured for body area network (BAN) operations in a limited multipath environment based on a constant symbol rate for BAN packet transmissions and based on M-ary PSK, differential M-ary PSK or rotated differential M-ary PSK modulation. The PHY layer is configured to transmit and receive data in a frequency band selected from the group consisting of: 402-405 MHz, 420-450 MHz, 863-870 MHz, 902-928 MHz, 950-956 MHz, 2360-2400 MHz, and 2400-2483.5 MHz.

CROSS-REFERENCE TO RELATED APPLICATION

The present application claims priority to: U.S. Provisional PatentApplication No. 61/169,048, filed on Apr. 14, 2009 (Attorney Docket No.TI-67981 PS); U.S. Provisional Patent Application No. 61/169,054, filedon Apr. 14, 2009 (Attorney Docket No. TI-67982PS); U.S. ProvisionalPatent Application No. 61/170,764, filed on Apr. 20, 2009 (AttorneyDocket No. TI-68012PS); U.S. Provisional Patent Application No.61/172,559, filed on Apr. 24, 2009 (Attorney Docket No. TI-68029PS);U.S. Provisional Patent Application No. 61/172,889, filed on Apr. 27,2009 (Attorney Docket No. TI-67981 PS1); U.S. Provisional PatentApplication No. 61/300,312, filed on Feb. 1, 2010 (Attorney Docket No.TI-69036PS); U.S. Provisional Patent Application No. 61/306,663, filedon Feb. 22, 2010 (Attorney Docket No. TI-69036PS1); U.S. ProvisionalPatent Application No. 61/313,440, filed on Mar. 12, 2010 (AttorneyDocket No. TI-67981PS2); U.S. Provisional Patent Application No.61/318,076, filed on Mar. 26, 2010 (Attorney Docket No. TI-68029PS.1);and U.S. Provisional Patent Application No. 61/319,063, filed on Mar.30, 2010 (Attorney Docket No. TI-6802PS2); all of which are herebyincorporated herein by reference.

This application also may contain subject matter that relates to thefollowing commonly assigned co-pending applications incorporated hereinby reference: “PHY Layer Options For Body Area Network (BAN) Devices,”U.S. Ser. No. 12/760,510, filed Apr. 14, 2010 (now U.S. Pat. No.8,275,342), Attorney Docket No. TI-67981; and “PHY Layer PPDUConstruction For Body Area Network (BAN) Devices,” U.S. Ser. No.12/760,513, filed Apr. 14, 2010 (now U.S. Pat. No. 8,391,228), AttorneyDocket No. TI-68012.

BACKGROUND

A medical body area network (BAN) refers to a low rate (e.g., less than1 Mbps), very low-power (e.g., less than 3 mA), very short-range (e.g.,less than 3 meters) wireless technology that is specifically designed tobe used in medical applications, such as digital band-aids andpacemakers. As an example, BAN technology could be implemented withdigital band-aids to measure vital statistics and wirelessly transferthe information to a larger network for further processing. Further, BANtechnology could be implemented with pacemakers to enable doctors tofine tune the device after implantation and to extract informationassociated with cardiac events. The implementation of a new wirelesstechnology, such as BAN, is not trivial.

SUMMARY

In at least some embodiments, a communication device includes atransceiver with a physical (PHY) layer. The PHY layer is configured forbody area network (BAN) operations in a limited multipath environmentbased on a constant symbol rate for BAN packet transmissions and basedon M-ary PSK, differential M-ary PSK or rotated differential M-ary PSKmodulation. The PHY layer is configured to transmit and receive data ina frequency band selected from the group consisting of: 402-405 MHz,420-450 MHz, 863-870 MHz, 902-928 MHz, 950-956 MHz, 2360-2400 MHz, and2400-2483.5 MHz.

BRIEF DESCRIPTION OF THE DRAWINGS

For a detailed description of exemplary embodiments of the invention,reference will now be made to the accompanying drawings in which:

FIG. 1 shows a Physical-Layer Protocol Data Unit (PPDU) in accordancewith embodiments of the disclosure;

FIGS. 2A-2G show tables of data-rate dependent parameter information fora Physical Layer Convergence Protocol (PLCP) header and Physical LayerData Unit (PSDU) in accordance with embodiments of the disclosure;

FIG. 3A-3B show tables of preamble sequence values in accordance withembodiments of the disclosure;

FIG. 4 shows a preamble structure in accordance with embodiments of thedisclosure;

FIG. 5 shows a block diagram of PLCP header construction in accordancewith embodiments of the disclosure;

FIG. 6 shows a PHY header format in accordance with embodiments of thedisclosure;

FIG. 7 shows a table of rate-dependent parameter information inaccordance with embodiments of the disclosure;

FIG. 8 shows a table of burst mode parameter information in accordancewith embodiments of the disclosure;

FIG. 9 shows a block diagram of a CRC-4 implementation in accordancewith embodiments of the disclosure;

FIG. 10 shows a block diagram of PSDU construction in accordance withembodiments of the disclosure;

FIG. 11 shows a block diagram of a side-stream scrambler in accordancewith the PSDU construction of FIG. 10;

FIG. 12 shows a scrambler seed selection table in accordance withembodiments of the disclosure;

FIG. 13A-13B show power spectral density charts for the PSDUconstruction of FIG. 10.

FIG. 14 shows an alternative PSDU construction in accordance withembodiments of the disclosure;

FIG. 15 shows a block diagram of a side-stream scrambler in accordancewith the PSDU construction of FIG. 14;

FIG. 16 shows a block diagram of PLCP header construction in accordancewith embodiments of the disclosure;

FIG. 17 shows a block diagram of an alternative PLCP header constructionin accordance with embodiments of the disclosure;

FIG. 18 shows a BCH encoding process for a single codeword in accordancewith embodiments of the disclosure;

FIGS. 19A and 19B show a spreading scheme in accordance with embodimentsof the disclosure;

FIG. 20 shows a table with GMSK symbol mapping information in accordancewith embodiments of the disclosure;

FIG. 21 shows a table with π/2-DBPSK mapping information in accordancewith embodiments of the disclosure;

FIG. 22 shows a table with π/4-DBPSK mapping information in accordancewith embodiments of the disclosure;

FIG. 23 shows a table with π/8-DBPSK mapping information in accordancewith embodiments of the disclosure;

FIG. 24 shows a table with center frequency and channel numberrelationship information in accordance with embodiments of thedisclosure;

FIG. 25 shows a table with channel number and preamble relationshipinformation in accordance with embodiments of the disclosure;

FIG. 26 shows a table with PHY layer timing parameter information inaccordance with embodiments of the disclosure;

FIG. 27 shows a table with inter-frame spacing parameter information inaccordance with embodiments of the disclosure;

FIG. 28 shows a table of channel bandwidth information as function offrequency band of operation in accordance with embodiments of thedisclosure;

FIG. 29 shows a transmit power-on ramp diagram in accordance withembodiments of the disclosure;

FIG. 30 shows a transmit power-down ramp diagram in accordance withembodiments of the disclosure;

FIG. 31 shows a table of permissible EVM information as a function ofconstellation size in accordance with embodiments of the disclosure;

FIG. 32 shows a table of receiver sensitivity information in accordancewith embodiments of the disclosure; and

FIG. 33 shows a system in accordance with embodiments of the disclosure.

DETAILED DESCRIPTION

The invention now will be described more fully hereinafter withreference to the accompanying drawings. This invention may, however, beembodied in many different forms and should not be construed as limitedto the embodiments set forth herein. Rather, these embodiments areprovided so that this disclosure will be thorough and complete, and willfully convey the scope of the invention to those skilled in the art. Oneskilled in the art may be able to use the various embodiments of theinvention.

Disclosed herein are options for a narrowband Physical (PHY) layer tosupport medical body area network (BAN) communications. The disclosedPHY layer options may be implemented by medical devices as well asdevices in communication with medical devices. At least some of thedisclosed PHY layer options may be adopted for the IEEE 802.15.6specification. Although designed for use with medical devices (e.g.,digital band-aids and pacemakers), it should be understood that thedisclosed PHY layer options are not limited to medical deviceembodiments. Rather, the disclosed PHY layer options enable a low rate(e.g., less than 1 Mbps), very low-power (e.g., less than 3 mA), veryshort-range (e.g., less than 3 meters) wireless technology that operatesin a limited multipath environment for use in any application.

In accordance with embodiments of the disclosure, the PHY is responsiblefor the following tasks: 1) activation and deactivation of the radiotransceiver; 2) clear channel assessment (CCA) and listen before talk(LBT) within the current channel; and 3) data transmission andreception. FIG. 1 shows a Physical-Layer Protocol Data Unit (PPDU) 100in accordance with embodiments of the disclosure. As shown, the PPDU 100comprises a Physical Layer Convergence Protocol (PLCP) preamble 102, aPLCP header 104, and a physical-layer service data unit (PSDU) 106.Disclosed herein are options for transforming the PSDU 106 into the PPDU100. Generally, at the transmitter-side, the PSDU 106 is pre-appendedduring transmission with the PLCP preamble 102 and the PLCP header 104in order to create the PPDU 100. At the receiver-side, the PLCP preamble102 and PLCP header 104 serve as aids in the demodulation, decoding anddelivery of the PSDU 106.

In FIG. 1, the PLCP header 104 is shown to be the second main componentof the PPDU 100. The purpose of PLCP header 104 is to convey informationabout PHY and MAC parameters to aid in decoding the PSDU 106 at thereceiver. In at least some embodiments, the PLCP header 106 comprises aPHY header field 112 (e.g., 15 bits in length), a header check sequence(HCS) field 114 (e.g., 4 bits in length), and a Bose, Ray-Chaudhuri,Hocquenghem (BCH) parity bits field (e.g., 12 bits in length). The BCHparity bits are added in order to improve the robustness of the PLCPheader 104. The PHY header field 112 may further be decomposed into aRATE field 120 (e.g., 3 bits in length), a LENGTH field (e.g., 8 bits inlength), a SCRAMBLER SEED field 128 (e.g., 1 bit in length), a BURSTMODE field 130 (e.g., 1 bit in length), and reserved bit fields 122 and126. The PLCP header 104 is transmitted using the given header data ratein the operating frequency band.

In FIG. 1, the PSDU 106 is shown to be the last component of the PPDU100. The PSDU 106 is formed by concatenating a MAC header 132 (e.g., 7bytes in length) with a MAC frame body 134 (e.g., 0-255 bytes in length)and a frame check sequence (FCS) (e.g., 2 bytes in length). In at leastsome embodiments, the PSDU 106 is scrambled and optionally encoded by aBCH code. The PSDU 106 may be transmitted using any of the availabledata rates available in the operating frequency band.

When transmitting the PPDU 100, the PLCP preamble 102 is sent first,followed by the PLCP header 104 and finally the PSDU 106. All multiplebyte fields are transmitted with least significant byte first and eachbyte is transmitted with the least significant bit (LSB) first. In atleast some embodiments, a compliant device is able to supporttransmission and reception in one of the following frequency bands:402-405 MHz, 420-450 MHz, 863-870 MHz, 902-928 MHz, 950-956 MHz,2360-2400 MHz and 2400-2483.5 MHz.

Various data-rate dependent parameters for each of these possiblefrequency bands of operation are provided herein and are intended toconform to established regulations, such as in the United States,Europe, Japan and Korea. FIGS. 2A-2G show tables with data-ratedependent parameter information for a PLCP header and a PSDU inaccordance with embodiments of the disclosure. The data-rate dependentparameters in FIGS. 2A-2F include parameters such as modulation type,symbol rate (in ksps), code rate (k/n), spreading factor (S),bandwidth-bit duration (BT), pulse shape, information data rate (inkbps), and whether support for a set of parameters is mandatory oroptional.

More specifically, the table 200 in FIG. 2A shows modulation parameterinformation for the frequency band 402-405 MHz, where a compliant devicesupports transmission and reception at a data rate of 75.9, 151.8 and303.6 kbps. The table 210 in FIG. 2B shows modulation parameterinformation for the frequency band 420-450 MHz, where a compliant devicesupports transmission and reception at a data rate of 75.9 and 151.8kbps. The table 220 in FIG. 2C shows modulation parameter informationfor the frequency band 863-870 MHz, where a compliant device supportstransmission and reception at a data rate of 101.2, 202.4 and 404.8kbps. The table 230 in FIG. 2D shows modulation parameter informationfor the frequency band 902-928 MHz, where a compliant device supportstransmission and reception at a data rate of 121.4, 242.9 and 485.7kbps. The table 240 in FIG. 2E shows modulation parameter informationfor the frequency band 950-956 MHz, where a compliant device supportstransmission and reception at a data rate of 101.2, 202.4 and 404.8kbps. The table 250 in FIG. 2F shows modulation parameter informationfor the frequency band 2360-2400 MHz, where a compliant device supportstransmission and reception at a data rate of 121.4, 242.9, 485.7 and971.4 kbps. The table 260 in FIG. 2G shows modulation parameterinformation for the frequency band 2400-2483.5 MHz, where a compliantdevice supports transmission and reception at a data rate of 121.4,242.9, 485.7 and 971.4 kbps.

In at least some embodiments, a packet-based time-division duplextechnique is implemented. In particular, one of the modes is based onrotated, differential M-PSK. In this modulation scheme, the informationis encoded in the phase difference between two consecutive symbols. Oneof the key components of the packet is the preamble, since it aids thereceiver in packet detection, acquisition, timing synchronization andcarrier-offset recovery. Disclosed herein are preamble optionsconstructed by concatenating a length-63 m-sequence with a 0101sequence.

FIG. 3A-3B show tables 300 and 310 with preamble sequence values inaccordance with embodiments of the disclosure. The tables 300 and 310show two different length-63 m-sequence preambles (bits b₀ to b₆₂), eachconcatenated with an extension sequence (bit b₆₃ to bit b₈₉). Eachm-sequence option is useful for packet detection, coarse-timingsynchronization and carrier-offset recovery. For example, a receiver mayimplement a simple correlator to search for the preamble. Thecorrelation is performed using the known transmitted preamble at thereceiver. When the preamble is on-air, the magnitude of the correlatorwill result in a peak when the known preamble matches the preambleon-air. This peak can be found using a simple hypothesis (i.e., whetherthe peak is greater than a predetermined threshold). If the peak isabove the threshold, then the receiver declares the packet to be on-air.In addition, the location of the peak provides information about thecoarse-timing synchronization. Finally, the value of the peak providesan estimate of the carrier-frequency offset. Since the bits aredifferentially encoded in the phase, the differential detection resultsin constant-phase offset in each symbol. So after implementing thecorrelator, the output is equal to the output of the correlator with azero frequency offset times a constant-phase offset. This constant-phaseoffset provide an estimate of the carrier-frequency offset, since thesymbol is known at the receiver.

Each extension sequence (bit b₆₃ to bit b₈₉) of tables 300 and 310 hastwo components: a timing sequence (b₆₃ to bit b₇₄) and a DC-balancedsequence (b₇₅ to bit b₈₉). In tables 300 and 310, the timing sequencecorresponds to a repeated 01 sequence and has various phase changes (orzero-crossings). These phase changes or zero-crossings can be exploitedto refine the receiver's estimate of the timing. The techniques forusing phase changes or zero-crossings to estimate and refining timingsynchronization are well known in the relevant art. Meanwhile, theDC-balanced sequence of tables 300 and 310 corresponds to a repeated 101sequence. The DC-balanced sequence provides enough state changes forreasonable clock recovery while achieving DC balance and boundeddisparity among adjacent data symbols

Wireless medical BAN devices are expected to be very low-power and verylow cost. This implies that the receiver may not have very strongfilters to reject adjacent channel interference (i.e., interference fromother BAN networks on either side of the current channel). Withoutstrong filters, energy from adjacent channels will bleed into thedesired channel. If a single preamble is defined within the system, thenit is possible that a preamble originating (being transmitted) on anadjacent channel will fold back into a receiver operating on the desiredchannel and will result in the packet detection algorithm declaring thata packet is on-air. Since this packet originated in an adjacent channel,this declaration is considered to be a false-alarm and will causevaluable energy to be wasted trying to decode the false packet.

By allocating two unique parameter sequences as in tables 300 and 310 inan intelligent manner, the occurrence of the false alarm conditionsdescribed above can be reduced. In at least some embodiments, preamblessuch as those given in tables 300 and 310 are defined in the table 2500of FIG. 25, where n, is the channel number and ranges from 0 to N−1 andwhere N is the total number of channels available. In at least someembodiments, the preambles will be transmitted at the symbol rate forthe desired band of operation and will be encoded using π/2-DBPSK. As analternative, FSK (frequency shift keying), GFSK (Gaussian frequencyshift keying) or GMSK (Gaussian minimum shift keying) may be usedinstead of π/2-DBPSK.

FIG. 4 shows a preamble structure 400 in accordance with embodiments ofthe disclosure. As shown, the preamble structure 400 comprises anm-sequence 402 and an extension sequence having a timing sequence 404and a DC-balanced sequence 406. In at least some embodiments, theextension sequence is constructed by concatenating the timing sequence404 with DC-balanced sequence 406. The timing sequence 404 maycorrespond to: K repetitions of any of (01) or (10), where K≧0. In someembodiments, the timing sequence 404 may be truncated. Meanwhile, theDC-balanced sequence 406 may correspond to: N repetitions of any of(100), (010), (001), (011), (101), (110), where N≧0. The DC-balancedsequence 406 also may be truncated. As an example, a length-14DC-balanced sequence may be generated by repeating any of (100), (010),(001), (011), (101), (110) 5 times; and taking the first 14 bits.Although the extension sequence of FIG. 4 shows the timing sequence 404before the DC-balanced sequence 406, alternative embodiments mayposition the DC-balanced sequence 406 before the timing sequence 404.

The advantage of the hybrid approach (concatenating the timing sequence404 with the DC-balanced sequence 406) is that the first portion of theextension sequence (the timing sequence 406) can be used to estimate andcorrect the fine timing and fine frequency offset, while the secondportion of the extension sequence (the DC-balanced sequence 404) can beused to estimate and remove any residual DC offset. The coarse timingand coarse frequency offset estimation can be performed on them-sequence portion of the preamble. Accordingly, the preamble structure400 with the extension sequence allows for a larger variety of receiverarchitectures. As an example, a 27-bit extension sequence may be: 01 0101 01 0 101 101 101 101 101 101 (with spaces provided for readability).This 27-bit extension sequence comprises a 9-bit timing sequence 404 anda 12-bit DC-balanced sequence 406. Another example is a 24-bit extensionsequence: 10 10 10 10 10 10 110 110 110 110. This 24-bit extensionsequence comprises a 12-bit timing sequence 404 and a 12-bit DC-balancedsequence 406. Another example is a 21-bit extension sequence: 10 10 1010 10 1 010 010 010 0. This 21-bit sequence comprises a 11-bit timingsequence 404 and an 10-bit DC-balanced sequence 406).

FIG. 5 shows a block diagram 500 of PLCP header (e.g., the PLCP header104) construction in accordance with embodiments of the disclosure. Asshown in the block diagram 500, a PHY header 502 is concatenated withHCS bits 506 and BCH parity bits 510. The PHY header 502, the HCS bits506, and the BCH parity bits 510 may correspond to the PHY header 112,the HCS field 114, and the BCH parity bit field 116 of FIG. 1. In FIG.5, the PHY header 502 is formed based on information provided by theMAC. At block 504, a 4-bit HCS value is calculated over the PHY header502 using the CRC-4 ITU polynomial: 1+x+x⁴ to generate the HCS field506. Finally, a BCH encoder 508 applies a BCH (31, 19) code 510, whichis shortened code derived from a BCH (63, 51) code, to the concatenationof the PHY header 502 (e.g., 15 bits in length) and the HCS field 506(e.g., 4 bits in length). Shortening is well known in the literature,and in this case, it involves appending the 19 information bits with 32zero bit in order to create the 51 bit message. After encoding, the 32zero bits are removed from the output bit stream. The resulting encodedbits are modulated using the appropriate parameters specified in FIGS.2A-2G for the desired frequency band of operation.

FIG. 6 shows a PHY header format 600 in accordance with embodiments ofthe disclosure. In general, the PHY header format 600 containsinformation about the data rate of the MAC frame body, the length of theMAC frame body (which does not include the MAC header or the FCS) andinformation about the next packet (e.g., whether it is being sent in aburst mode, where multiple packets are transmitted consecutively using aminimum inter-frame spacing). More specifically, the PHY header format600 of FIG. 6 comprises 15 bits, numbered from 0 to 14 as illustrated,LBS to MSB. Bits 0-2 correspond to a RATE field, which conveysinformation about the type of modulation, the symbol rate, frequencydeviation or pulse shape, the coding rate, and the spreading factor usedto transmit the PSDU. Bits 4-11 correspond to a LENGTH field, with theleast-significant bit being transmitted first. In at least someembodiments, the LENGTH field corresponds to an unsigned 8-bit integerthat indicates the number of un-coded information bytes in the MAC framebody (which does not include the MAC header or the FCS). Bit 13corresponds to a burst mode bit that indicates whether or not the packetis being transmitted in the burst (streaming) mode. Bit 14 correspondsto a scrambler seed bit that encodes a scrambler seed. In at least someembodiments, the MAC sets the scrambler seed bit (SS) according to aseed value chosen for the data scrambler later described. All other bits(e.g., bits 3 and 12) of the PHY header format 600 are reserved forfuture use and are set to zero.

FIG. 7 shows a table 700 of rate-dependent parameter information inaccordance with embodiments of the disclosure. In table 700, theparameters vary based on the value of the bits in the RATE field (bitsR0-R2) of the PHY header format 600. As shown, the data rates related toeach of RATE field values 000, 100, 010, and 110 varies depending on thefrequency band of operations. Further, certain data rates are reservedin table 700.

FIG. 8 shows a table 800 of burst mode parameter information inaccordance with embodiments of the disclosure. In at least someembodiments, the MAC sets the burst mode (BM) bit to indicate whetherthe next packet is part of a packet “burst” (i.e., burst modetransmission). In burst mode, the inter-frame spacing is equal to thepMIFS value (20 μs) shown in table 2600 of FIG. 26.

FIG. 9 shows a block diagram 900 of a CRC-4 implementation in accordancewith embodiments of the disclosure. In accordance with at least someembodiments, the PHY header (e.g., PHY header 112 or 502) is protectedwith a 4-bit (CRC-4 ITU) header check sequence (HCS). The HCScorresponds to the ones complement of the remainder generated by themodulo-2 division of the PHY header by the polynomial: 1+x+x⁴. The HCSbits are processed in the transmit order. A schematic of the processingorder is shown in block diagram 900. The registers for the CRC-4operation are initialized to all ones, and the output is the onescompliment of the shift-register values.

FIG. 10 shows a block diagram 1000 of PSDU construction in accordancewith embodiments of the disclosure. During PSDU construction, theconcatenate block 1002 forms the non-scrambled PSDU by pre-pending the7-byte MAC header to the MAC frame body and appending a 2-byte FCS tothe result. If the code rate (k/n)<1, the resulting PSDU is divided intoblocks of messages, where each message may contain shortened bitsinserted by the insert shortened bits block 1006. The operation of theinsert shortened bits block 1006 is based in part on the input receivedfrom BCH encoding algorithm 1004. The resulting messages are thenencoded into codewords using a BCH encoder 1008 to achieve the desiredcode rate. In at least some embodiments, the BCH encoder 1008 supports acode rate of 51/63. Finally, the shortened bits are removed from each ofthe codewords by the remove shortened bits block 1010. Pad bits are thenadded by the add pad bits block 1012 in order to ensure alignment on asymbol boundary. If the spreading factor is 2 or 4, the resultingun-coded or coded bits are spread by spreader 1014 using a repetitioncode, and then interleaved using bit interleaver 1016. The resulting bitstream is scrambled by scrambler 1018. The resulting bit steam from thescrambler 1018 is then mapped onto the appropriate constellation bysymbol mapper 1020 based on the data rate and frequency band ofoperation. In block diagram 1000, the scrambler 1018 is located afterthe bit interleaver 1016 and runs at the (symbol rate)×(the number ofbits per symbol). This makes the transmit signal more random and removesany spectral lines in the power spectral density of the transmit signal.

In alternative embodiments, the scrambler 1018 is positioned between theconcatenate block 1002 and the insert shortened bits block 1006. In suchembodiments, the PSDU output from the concatenate block 1002 isscrambled and is then processed by the insert shortened bits block 1006,the BCH encoder 1008, the remove shortened bits block 1010, the add padbits block 1012, the spreader 1014 and the bit interleaver 1016 asdescribed for FIG. 10. In other words, the position of the scramblingstep for PSDU construction may vary.

FIG. 11 shows a block diagram 1100 of a side-stream scrambler inaccordance with embodiments of the disclosure. The block diagram 1100corresponds, for example, to the operation of scrambler 1018 in FIG. 10.In block diagram 1100, a side-stream scrambler with polynomialG(x)=1+x²+x¹²+x¹³+x¹⁴ is used to whiten a PSDU. The output of thescrambler is generated as: x[n]=x[n−2]{circle around (+)}x[n−12]{circlearound (+)}x[n−13]{circle around (+)}x[n−14], where “{circle around(+)}” denotes modulo-2 addition. The table 1200 of FIG. 12 defines theinitialization vector, x_(init), for the side-stream scrambler as afunction of the scrambler seed (SS) value. In at least some embodiments,the MAC sets the scrambler seed to 0 when the PHY is initialized and thescrambler seed is incremented, using a 1-bit rollover counter, for eachframe sent by the PHY. At the receiver-side, the side-streamde-scrambler is initialized with the same initialization vector,x_(init), used by the transmitter. The initialization vector isdetermined from the scrambler seed value in the PHY header of thereceived frame.

FIGS. 13A-13B show power spectral density charts 1300 and 1310 for thePSDU construction of FIG. 10. More specifically, the chart 1300corresponds to a power spectral density when the spreading factor forthe PSDU construction of FIG. 10 is two. Meanwhile, the chart 1310corresponds to a power spectral density when the spreading factor forthe PSDU construction of FIG. 10 is four. Importantly, spectral linesare reduced or eliminated for the spectral density charts 1300 and 1310by positioning spreader 1014 before bit interleaver 1016.

FIG. 14 shows a block diagram 1400 of an alternative PSDU constructionin accordance with embodiments of the disclosure. As shown, for the PSDUconstruction of block diagram 1400, the scrambler 1420 is located afterthe symbol mapper 1418 and runs at symbol-level (in symbol rate). ThePSDU construction components for block diagram 1400 are similar to thecorresponding PSDU construction components for block diagram 1300,except for the scrambler 1418.

FIG. 15 shows a block diagram of a side-stream scrambler (e.g.,scrambler 1418) in accordance with the PSDU construction of FIG. 14. Thescrambler 1418 has the same polynomial G(x)=1+x²+x¹²+x¹³+x¹⁴ as in thePSDU. The scrambler 1418 multiples the symbols (generallycomplex-numbered) by the scrambling sequence after mapping the binary{0,1} sequence to {+1,−1} sequence. Note that the mapping can either bebit 0 to −1, bit 1 to +1; or bit 0 to +1, bit 1 to −1, as long as themapping is consistent between a transmitter and receiver.

FIG. 16 shows a block diagram 1600 of PLCP header construction inaccordance with embodiments of the disclosure. As shown, the blockdiagram 1600 comprises a concatenate block 1602 that receives a PHYheader and HCS as input. A BCH encoder 1604 operates on the output ofthe concatenate block 1602, followed by a spreader 1606, a bitinterleaver 1608, a scrambler 1610, and a symbol mapper 1612. In thePLCP header construction of block diagram 1600, the scrambler 1610 islocated after the bit interleaver 1608 and before the symbol mapper1612. In at least some embodiments, the scrambler 1610 corresponds tothe scrambler 1018 shown for the PSDU construction block diagram 1000 ofFIG. 10.

In the PLCP header construction of block diagram 1600, the scramblingoperation (at symbol level) of the scrambler 1610 is applied to PLCPheader and the initial seed for the scrambler 1610 is known a priori tothe receiver. One possible method for pre-assigning the scrambler seedis to map the even channels to scrambler seed 0, and odd channels toscrambler seed 1, or vice versa. As an example, if devices are operatingon channel 2 (an even channel), then scrambler seed 0 would be used toscramble the PLCP header. At the end of the PLCP header, the scrambler1610 would be re-initialized with the scrambler seed specified by theMAC (either scrambler seed 0 or 1), and the re-initialized scramblerwould be used to scramble the PSDU. If devices are operating on channel3 (an odd channel), then scrambler seed 1 would be used to scrambler thePLCP header. Again, at the end of the PLCP header, the scrambler 1610would be re-initialized with the scrambler seed specified by the MAC(either scrambler seed 0 or 1), and the re-initialized scrambler wouldbe used to scramble the PSDU. Although only one method is shown forpre-assigning the scrambler seed for the PLCP header based on thechannel information, there are many other ways to pre-assign thescrambler seed for the PLCP header.

FIG. 17 shows a block diagram 1700 of an alternative PLCP headerconstruction in accordance with embodiments of the disclosure. The PLCPheader construction components for block diagram 1700 are similar to thecorresponding PLCP header construction components for block diagram1600, except for the scrambler 1712. In the PLCP header construction ofblock diagram 1700, the scrambler 1712 is located after the symbolmapper 1710 and may correspond to scrambler 1420 for PSDU constructionblock diagram 1400.

FIG. 18 shows a BCH encoding process 1800 for a single codeword inaccordance with embodiments of the disclosure. The process 1800 startswith message bits at block 1802. At block 1804, shortened bits are addedto the message bits. At block 1806, parity bits are added to the messagebits and shortened bits. Finally, at block 1808, the shortened bits areremoved, while the message bits and parity bits remain.

The BCH encoding process may be performed by any of the BCH encodersmentioned herein (e.g., BCH encoders 1008, 1408, 1604, 1704), which mayrepresent the same BCH encoder. The scrambled or non-scrambled PSDU isencoded by computing the number of bits in the PSDU. In at least someembodiments, the number of bits in a PSDU is calculated asN_(PSDU)=(N_(MACheader)+N_(MACFrameBody)+N_(FCS))×8, where N_(MACheader)is the number of bytes in the MAC header, N_(MACFrameBody) is the numberof bytes in the MAC frame body and N_(FCS) is the number of bytes in theFCS. The number of BCH codewords is then calculated as

${N_{CW} = \left\lceil \frac{N_{PSDU}}{k} \right\rceil},$

where k is the number of message bits for the selected BCH code. Thenumber of shortening bits, N_(shorten), to be padded to the N_(PSDU)data bits before encoding is computed as N_(shorten)=N_(CW)×k−N_(PSDU).The shortening bits are equally distributed over all N_(CW) codewordswith the first rem(N_(shorten),N_(CW)) codewords being shortened one bitmore than the remaining codewords. Assuming

${N_{spcw} = \left\lfloor \frac{N_{shorten}}{N_{CW}} \right\rfloor},$

the first rem(N_(shorten),N_(CW)) codewords will have N_(spcw)+1shortened bits (message bits that are set to 0), while the remainingcodewords will have N_(spcw) shortened bits. After encoding, theshortened bits are discarded prior to transmission (i.e., the shortenedbits are never transmitted on-air).

For a BCH (63, 51) encoder, the generator polynomial for a systematicBCH (63, 51, t=2) code is given as g(x)=1+x³+x⁴+x⁵+x⁸+x¹⁰+x^(12.) Theparity bits are determined by computing the remainder polynomial as

${{r(x)} = {{\sum\limits_{i = 0}^{11}\; {r_{i}x^{i}}} = {x^{12}{m(x)}{mod}\; {g(x)}}}},$

where m(x) is the message polynomial

${{m(x)} = {{\sum\limits_{i = 0}^{50}\; {m_{i}x^{i}\mspace{14mu} {and}\mspace{14mu} r_{i}i}} = 0}},\ldots \mspace{14mu},11.$

Further, m_(i),i=0, . . . ,50 are elements of GF(2). The messagepolynomial m(x) is created as follows: m₅₀ is the first bit of themessage and m₀ is the last bit of the message, which may be a shortenedbit. The order of the parity bits is as follows: r₁₁ is the first paritybit transmitted, r₁₀ is the second parity bit transmitted, and r₀ is thelast parity bit transmitted.

Pad bits are appended after the BCH encoder to ensure that the bitstream aligns on a symbol boundary. The number of pad bits, N_(pad),that are inserted is a function of the number of PSDU bits N_(PSDU), thenumber of codewords N_(CW), the number of parity bits (n−k), and themodulation constellation size M determined from

$N_{pad} = {{{\log_{2}(M)} \times \left\lceil \frac{N_{PSDU} + {N_{CW} \times \left( {n - k} \right)}}{\log_{2}(M)} \right\rceil} - {\left\lbrack {N_{PSDU} + {N_{CW} \times \left( {n - k} \right)}} \right\rbrack.}}$

The pad bits are appended to the scrambled and encoded PSDU, where allof the appended pad bits are set to 0. In the case of un-codedtransmission, N_(CW) is set to zero.

FIGS. 19A and 19B show a spreading scheme 1900 in accordance withembodiments of the disclosure. As shown in spreading scheme 1900, for aspreading factor of 2, each input bit is repeated two times. For aspreading factor of 4, each input bit is repeated four times.

In the PSDU construction of block diagrams 1000 (FIGS. 10) and 1400(FIG. 14), the bits interleavers 1016 and 1416 perform bit interleavingas given below. The same or similar bit interleaving process also may beperformed by the bit interleavers 1608 and 1708 of the respective PLCPconstruction block diagrams 1600 (FIGS. 16) and 1700 (FIG. 17). Althoughnot required, the bit interleavers 1016, 1416, 1608, and 1708 mayrepresent a single bit interleaver.

In at least some embodiments, the spreader output (e.g., from spreader1014, 1414, 1606, 1706) is interleaved by a bit interleaver prior tomodulation to provide robustness against error propagation. The exactstructure of the bit interleaver depends on the number of un-coded orcoded bits that will be transmitted on-air, which is given asN_(total)=N_(PSDU)+N_(CW)×(n−k)+N_(pad), where N_(CW) is set to zero inthe case of un-coded transmission. If rem(N_(total),2)=0, the bitinterleaving operation is performed by first grouping the spread bitsinto blocks of 2S bits, where S is the spreading factor, and then usinga block interleaver of size S×2 to permute the bits. Using sequencesa(i) and b(i) (where i=0, 1, . . . , 2S−1) to respectively represent theinput and output bits of the S×2 bit interleaver, the output of an S×2bit interleaver is given as

${{b(i)} = {{{a\left\lbrack {{S \times {{rem}\left( {i,2} \right)}} + \left\lfloor \frac{i}{2} \right\rfloor} \right\rbrack}\mspace{14mu} i} = 0}},1,\ldots \mspace{14mu},{{2\; S} - 1.}$

If rem(N_(total),2)=1, the bit interleaving operation is performed bygrouping the first 3S spread bits into a single block and then using ablock interleaver of size S×3 to permute the bits within that singleblock. Using sequences a(i) and b(i) (where i=0, 1, . . . , 3S−1) torespectively represent the input and output bits of a S×3 bitinterleaver, the output of the S×3 bit interleaver is given as

${{b(i)} = {{{a\left\lbrack {{S \times {{rem}\left( {i,3} \right)}} + \left\lfloor \frac{i}{3} \right\rfloor} \right\rbrack}\mspace{14mu} i} = 0}},1,\ldots \mspace{14mu},{{3\; S} - 1.}$

The remaining spread bits are then grouped into blocks of 2S bits andinterleaved using a block interleaver of size S×2.

In the PSDU construction of block diagrams 1000 (FIGS. 10) and 1400(FIG. 14), the symbol mappers 1020 and 1418 perform may perform GMSKsymbol mapping. The same or similar GMSK symbol mapping also may beperformed by the symbol mappers 1612 and 1710 of the respective PLCPconstruction block diagrams 1600 (FIG. 16) and 1700 (FIG. 17). Althoughnot required, the symbols mappers 1020, 1418, 1612, and 1710 mayrepresent a single bit interleaver.

FIG. 20 shows a table 2000 with GMSK symbol mapping information inaccordance with embodiments of the disclosure. For the GMSKconstellation, the un-coded or coded, potentially spread and interleavedbinary bit stream b(n), n=0, 1, . . . , N−1 is mapped onto acorresponding frequency deviation Δf, which is the product of the symbolrate and a modulation index of 0.5. The relationship between the bitstream b(n) and the frequency deviation is given in table 2000.

For the D-PSK constellations, the coded potentially spread andinterleaved bit stream is mapped onto one of three rotated anddifferentially-encoded constellations: π/2-DBPSK, π/4-DQPSK, orπ/8-D8PSK. The encoded information is carried in the phase transitionsbetween symbols. For the PLCP preamble to PLCP header transition, thephase change is relative to the last symbol for the PLCP preamble. Forthe PLCP header to PSDU transition, the phase change is relative to thelast symbol for the PLCP header. The binary bit stream b(n), n=0, 1, . .. , N−1 is mapped onto a corresponding complex-values sequence S(k),k=0,1, . . . , (N/log₂(M))−1 as S(k)=S(k−1)exp(jφ_(k)) k=1, 2, . . . ,(N/log₂(M))−1, where S(0)=exp(jπ/M) and the relationship between the bitstream b(n) and the phase change φ_(k) is given in the tables 2100,2200, 2300 of FIGS. 21-23 for π/2-DBPSK (M=2), π/4-DQPSK (M=4), orπ/8-D8PSK (M=8), respectively.

As previously mentioned, a compliant device is able to supporttransmission and reception in one of the following frequency bands:402-405 MHz, 420-450 MHz, 863-870 MHz, 902-928 MHz, 950-956 MHz,2360-2400 MHz and 2400-2483.5 MHz. FIG. 24 shows a table 2400 withcenter frequency and channel number relationship information inaccordance with embodiments of the disclosure. The mapping functionsg₁(n_(c)) and g₂(n_(c)) used in the 420-450 MHz and 863-870 MHzfrequency bands are respectively defined as

${g_{1}\left( n_{c} \right)} = \left\{ {\begin{matrix}n_{c} & {0 \leq n_{c} \leq 1} \\{n_{c} + 6.875} & {2 \leq n_{c} \leq 4} \\{n_{c} + 13.4} & {n_{c} = 5} \\{n_{c} + 35.025} & {6 \leq n_{c} \leq 7} \\{n_{c} + 40.925} & {8 \leq n_{c} \leq 9} \\{n_{c} + 47.25} & {10 \leq n_{c} \leq 11}\end{matrix},{{{and}{g_{2}\left( n_{c} \right)}} = \left\{ {\begin{matrix}n_{c} & {0 \leq n_{c} \leq 7} \\{n_{c} + 0.5} & {n_{c} = 8} \\{n_{c} + 1} & {9 \leq n_{c} \leq 12} \\{n_{c} + 1.5} & {n_{c} = 13}\end{matrix}.} \right.}} \right.$

FIG. 25 shows a table 2500 with channel number and preamble relationshipinformation in accordance with embodiments of the disclosure. In table2500, the allocation of preambles (e.g., the preamble sequences shownfor FIGS. 3A-3C) is defined to reduce occurrence of false alarmconditions, where n_(c) is the channel number and ranges from 0 to N−1and where N is the total number of channels available.

FIG. 26 shows a table 2600 with PHY layer timing parameter informationin accordance with embodiments of the disclosure. The values for pEDTimeand pCCATime are either those specified in table 2600 or the valuesspecified by the local regulatory requirements (e.g., whichever islower).

FIG. 27 shows a table 2700 with inter-frame spacing parameterinformation in accordance with embodiments of the disclosure. As shown,the parameter SIFS has a value equal to pSIFS and the parameter MIFS hasa value equal to pMIFS. In at least some embodiments, pSIFS isapproximately 50 μs and pMIFS is approximately 20 μs as shown in table2600 of FIG. 26. Other parameters are also related to the pSIFS andpMIFS values. For example, the Receive-to-Transmit (RX-to-TX) turnaroundtime is equal to or less than pSIFS. The RX-to-TX turnaround time isdefined as the time elapsed from when the last sample of the lastreceived symbol is present on the air interface, to the time when firstsample of the first transmitted symbol of the PLCP preamble for the nextframe is present on the air interface. Further, the Transmit-to-Receive(TX-to-RX) turnaround time is equal to or less than pSIFS. The TX-to-RXturnaround time is defined as the time elapsed from when the last sampleof the last transmitted symbol is present on the air interface until thetime when the receiver is ready to begin the reception of first samplefor the next PHY frame. Further, for burst mode transmissions, theinter-frame spacing between uninterrupted successive transmissions by adevice shall be fixed to pMIFS. The inter-frame spacing is defined asthe time elapsed from when the last sample of the last transmittedsymbol is present on the air interface, to the time when the firstsample of the first transmitted symbol of the PLCP preamble for thefollowing packet is present on the air interface. Further, the centerfrequency switch time is defined as the interval from when the PHYtransmits or receives the last valid symbol on one center frequencyuntil it is ready to transmit or receive the next symbol on a differentcenter frequency. In at least some embodiments, the center frequencyswitch time does not exceed the pChannelSwitchTime value in table 2600(i.e., 100 μs).

Various transmitter parameters are disclosed herein. FIG. 28 shows atable 2800 with channel bandwidth information as function of frequencyband of operation in accordance with embodiments of the disclosure. Inat least some embodiments, the transmitted spectral mask shall be lessthan −X dBr (dB relative to the maximum spectral density of the signal)for |f−f_(c)|≧f_(BW)/2, where f_(c) is channel center frequency andf_(BW) is the channel bandwidth and is a function of the frequency bandof operation. Values for f_(BW) in relation to the frequency band ofoperation are given in table 2800 of FIG. 28.

In accordance with embodiments, the transmitted spectral density shouldcomply with all regulations defined by local regulatory bodies. Further,a transmitter should be capable of transmitting at least −10 dBm EIRP inall frequency bands, except for 402-405 MHz, where a transmitter shallbe capable of transmitting at most −16 dBm EIRP. Devices should transmitlower power when possible in order to reduce interference to otherdevices and systems. Again, the maximum transmit power is limited bylocal regulatory bodies.

FIG. 29 shows a transmit power-on ramp diagram 2900 in accordance withembodiments of the disclosure. As shown in diagram 2900, the transmitpower-on ramp for 10% to 90% of maximum power is not more than 5symbols. Similarly, FIG. 30 shows a transmit power-down ramp diagram3000 in accordance with embodiments of the disclosure. As shown in thediagram 3000, the transmit power-down ramp for 90% to 10% maximum poweris not more than 5 symbols. As needed, the transmit power ramps areconstructed such that the emissions conform to the local spuriousfrequency regulations.

Further, in at least some embodiments, the transmitted center frequencytolerance is ±20 ppm maximum. The symbol clock frequency tolerance is±20 ppm maximum. With regard to clock synchronization, the transmitcenter frequencies and the symbol clock frequency are derived from thesame reference oscillator.

The modulation accuracy of the transmitter is determined via anerror-vector magnitude (EVM) measurement, which is calculated over Nbaud-spaced received complex values (Î_(k), {circumflex over (Q)}_(k)).A decision is made for each received complex value. The ideal positionof the chosen symbol is represented by the vector (I_(k), Q_(k)). Theerror vector (δI_(k), δQ_(k)) is defined as the distance from the idealposition to the actual position of the received complex values, i.e.,(Î_(k), {circumflex over (Q)}_(k))=(I_(k), Q_(k))+(δI_(k), δQ_(k)).

The EVM is defined as shown in the equation below:

${{EVM} = {\sqrt{\frac{\frac{1}{N}{\sum\limits_{k = 1}^{N}\; \left( {{\delta \; I_{k}^{2}} + {\delta \; Q_{k}^{2}}} \right)}}{S^{2}}} \times 100\%}},$

where S is the magnitude of the vector to the ideal constellation point.A transmitter shall have EVM values less than or equal to those listedin the table 3100 of FIG. 31, where the measure for N=TBD (to bedetermined) symbols. In at least some embodiments, the EVM is measuredon baseband I and Q samples after the received signal is passed througha reference receiver, which shall perform the following operations:matched SRRC filtering, carrier-frequency offset estimation and symboltiming recovery while making the measurements.

Various receiver parameters are disclosed herein. FIG. 32 shows a table3200 for receiver sensitivity information in accordance with embodimentsof the disclosure. For a packet error rate (PER) of less than 10% with aPSDU of 255 bytes, the minimum receiver sensitivity numbers in AWGN forthe highest data rate in each operating frequency band are listed in thetable 3200. The minimum input levels are measured at the antennaconnector, where a noise figure of 10 dB (referenced at the antenna), animplementation loss of 6 dB, and antenna gain of 0 dBi for both thetransmitter and receiver are assumed.

In at least some embodiments, the receiver energy detection (ED)measurement is an estimate of the received signal power within thebandwidth of the channel. It is intended for use by a network/MAC layeras part of a channel selection algorithm. No attempt is made to identifyor decode signals on the channel. Further the minimum ED value (zero)indicates received power less than either 10 dB above the specifiedreceiver sensitivity (see table 3200 of FIG. 32), or a value prescribedlocal regulatory requirements (e.g., whichever is lower). In at leastsome embodiments, the range of received power spanned by the ED valuesis at least 40 dB. Within this range, the mapping from the receivedpower in decibels to ED value is linear with an accuracy of ±6 dB.Further, the ED measurement time, to average over, is either pEDTime (8preamble symbol periods) or a value prescribed by local regulatoryrequirements (e.g., whichever is longer in duration).

In at least some embodiments, the PHY is able to perform CCA accordingto at least one of the following three methods: CCA mode 1, CCA mode 2,and CCA mode 3. CCA mode 1 corresponds to an “energy above threshold”mode, in which CCA reports a busy medium upon detecting any energy abovethe ED threshold. CCA mode 2 corresponds to a “carrier sense only” mode,in which CCA reports a busy medium only upon the detection of a signalcompliant with this standard with the same modulation andcharacteristics of the PHY that is currently in use by the device. Thissignal may be above or below the ED threshold. The CCA detection time isequal to pCCATime. The CCA mode 3 corresponds to a “carrier sense withenergy above threshold” mode, in which CCA reports a busy medium using alogical combination of: 1) detection of a signal with the modulation andcharacteristics of this standard; and 2) energy above the ED threshold,where the logical operator may be “AND” or “OR”. The CCA parameters aresubject to the following criteria: 1) the ED threshold corresponds to areceived signal power as prescribed in table 3200 of FIG. 32; and 2) theCCA detection time is equal to pCCATime (see table 2600 of FIG. 26). AnyCCA procedures required by local regulatory requirements should also besupported.

FIG. 33 shows a system 3300 in accordance with embodiments of thedisclosure. The system 3300 comprises BAN devices 3302 and 3312 thatcommunicate with each other based on implementation of at least some ofthe PHY options disclosed herein. The system could alternative comprisesadditional devices in communication with each other. The BAN devices3302 and 3312 may correspond to medical devices (e.g., digital band-aidsand pacemakers), but are not limited thereto. As shown, the BAN device3302 comprises a transceiver 3304 with BAN PHY layer 3306. The BAN PHYlayer 3306 implements at least some of the PHY options disclosed herein.Similarly, the BAN device 3312 comprises a transceiver 3314 with BAN PHYlayer 3316, where the BAN PHY layer 3316 implements at least some of thePHY options disclosed herein. With the disclosed PHY options, the BANdevices 3302 and 3312 are each able to: 1) activate and deactivate aradio transceiver; 2) perform CCA within the current channel; and 3)transmit and receive data. Although BAN technology was developed formedical uses, the PHY options are not necessarily limited to aparticular field. Again, the disclosed PHY layer options enable a lowrate (e.g., less than 1 Mbps), very low-power (e.g., less than 3 mA),very short-range (e.g., less than 3 meters) wireless technology for usein any application.

In at least some embodiments, the PHY layer operations of PHY layers3306 and 3316 may be implemented in hardware such as an applicationspecific integrated circuit (ASIC). Additionally or alternatively, atleast some of the PHY layer operations described herein are implementedby a processor that executes software. Additionally or alternatively,specialized hardware accelerators may be implemented to perform at leastsome of the PHY operations described herein.

The above discussion is meant to be illustrative of the principles andvarious embodiments of the present invention. Numerous variations andmodifications will become apparent to those skilled in the art once theabove disclosure is fully appreciated. For example, ordinary D-BPSK,D-QPSK, or D-8PSK may be used for BAN devices instead of the π/M shiftedversions disclosed herein. Further, ordinary BPSK, QPSK and 8PSK may beused for BAN devices. It is intended that the following claims beinterpreted to embrace all such variations and modifications.

What is claimed is:
 1. A communication device, comprising: a transceiverwith a physical (PHY) layer, wherein the PHY layer is configured forbody area network (BAN) operations in a limited multipath environmentbased on a constant symbol rate for BAN packet transmissions and basedon M-ary PSK, differential M-ary PSK or rotated differential M-ary PSKmodulation, wherein the PHY layer is configured to transmit and receivedata in a frequency band selected from the group consisting of: 402-405MHz, 420-450 MHz, 863-870 MHz, 902-928 MHz, 950-956 MHz, 2360-2400 MHz,and 2400-2483.5 MHz.
 2. The communication device of claim 1 wherein, ifthe PHY layer uses the 402-405 MHz frequency band, a 000 rate fieldvalue in a PHY header indicates a rate of 75.9 kbps, a 100 rate fieldvalue in the PHY header indicates a rate of 151.8 kbps, a 010 rate fieldvalue in the PHY header indicates a rate of 303.6 kbps, and a 110 ratefield value in the PHY header indicates a rate of 455.4 kbps.
 3. Thecommunication device of claim 2 wherein, if the PHY layer uses the402-405 MHz frequency band, a transmitted spectral mask shall be lessthan −20 −X dBr for |f−f_(c)|≧f_(BW)/2, where f_(c) is channel centerfrequency and f_(BW) is 300 kHz.
 4. The communication device of claim 1wherein, if the PHY layer uses the 420-450 MHz frequency band, a 000rate field value in a PHY header indicates a rate of 75.9 kbps, a 100rate field value in the PHY header indicates a rate of 151.8 kbps, a 010rate field value in the PHY header indicates a rate of 187.5 kbps. 5.The communication device of claim 4 wherein, if the PHY layer uses the420-450 MHz frequency band, a transmitted spectral mask shall be lessthan −40−X dBr for |f−f_(c)|÷f_(BW)/2, where f_(c) is channel centerfrequency and f_(BW) is 320 kHz.
 6. The communication device of claim 1wherein, if the PHY layer uses the 863-870 MHz frequency band or the950-956 MHz frequency band, a 000 rate field value in a PHY headerindicates a rate of 101.2 kbps, a 100 rate field value in the PHY headerindicates a rate of 202.4 kbps, a 010 rate field value in the PHY headerindicates a rate of 404.8 kbps, and a 110 rate field value in the PHYheader indicates a rate of 607.1 kbps.
 7. The communication device ofclaim 6 wherein, if the PHY layer uses the 863-870 MHz frequency band orthe 950-956 MHz frequency band, a transmitted spectral mask shall beless than −20−X dBr for |f−f_(c)|≧f_(BW)/2, where f_(c) is channelcenter frequency and f_(BW) is 400 kHz.
 8. The communication device ofclaim 1 wherein, if the PHY layer uses the 902-928 MHz frequency band, a000 rate field value in a PHY header indicates a rate of 121.4 kbps, a100 rate field value in the PHY header indicates a rate of 242.9 kbps, a010 rate field value in the PHY header indicates a rate of 485.7 kbps,and a 110 rate field value in the PHY header indicates a rate of 728.6kbps.
 9. The communication device of claim 8 wherein, if the PHY layeruses the 902-928 MHz frequency band, a transmitted spectral mask shallbe less than −20−X dBr for |f−f_(c)|≧f_(BW)/2, where f_(c) is channelcenter frequency and f_(BW) is 500 kHz.
 10. The communication device ofclaim 1 wherein, if the PHY layer uses the 2360-2400 MHz frequency bandor the 2400-2483.5 MHz frequency band, a 000 rate field value in a PHYheader indicates a rate of 121.4 kbps, a 100 rate field value in the PHYheader indicates a rate of 242.9 kbps, a 010 rate field value in the PHYheader indicates a rate of 485.7 kbps, and a 110 rate field value in thePHY header indicates a rate of 971.4 kbps.
 11. The communication deviceof claim 6 wherein, if the PHY layer uses the 2360-2400 MHz frequencyband or the 2400-2483.5 MHz frequency band, a transmitted spectral maskshall be less than −20−X dBr for |f−f_(c)|≧f_(BW)/2, where f_(c) ischannel center frequency and f_(BW) is 1 MHz.
 12. The communicationdevice of claim 1 wherein the PHY layer uses a pSIFS value ofapproximately 50 μs and a pMIFS of approximately 20 μs.
 13. Thecommunication device of claim 12 wherein the PHY layer uses a SIFS valueequal to pSIFS and a MIFS value equal to pMIFS.
 14. The communicationdevice of claim 1 wherein the PHY layer uses a pEDTime of 8 preamblesymbols.
 15. The communication device of claim 1 wherein the PHY layeruses a pCCATime of 63 preamble symbols.
 16. The communication device ofclaim 1 wherein the PHY layer uses a pChannelSwitchTime of 100 μs. 17.The communication device of claim 1 wherein the PHY layer comprises atransmitter capable of transmitting at least −10 dBm EIRP for the420-450 MHz, 863-870 MHz, 902-928 MHz, 950-956 MHz, 2360-2400 MHz, and2400-2483.5 MHz frequency bands, and wherein the transmitter is capableof transmitting at most −16 dBm EIRP for the 402-405 MHz frequency band.18. The communication device of claim 1 wherein a transmit power-on rampfor the communication device changes from 10% to 90% of maximum power in5 symbols or less.
 19. The communication device of claim 1 wherein atransmit power-down ramp for the communication device changes from 90%to 10% of maximum power in 5 symbols or less.
 20. The communicationdevice of claim 1 wherein a transmitted center frequency tolerance forthe PHY layer is a maximum of ±20 ppm.
 21. The communication device ofclaim 20 wherein a symbol clock frequency tolerance for the PHY layer isa maximum of ±20 ppm.
 22. The communication device of claim 21 whereintransmit center frequencies and a symbol clock frequency for the PHYlayer are derived from a single reference oscillator.
 23. Thecommunication device of claim 1 wherein modulation accuracy isdetermined as${{EVM} = {\sqrt{\frac{\frac{1}{N}{\sum\limits_{k = 1}^{N}\; \left( {{\delta \; I_{k}^{2}} + {\delta \; Q_{k}^{2}}} \right)}}{S^{2}}} \times 100\%}},$computed over N baud-spaced received complex values (Î_(k), {circumflexover (Q)}_(k)), where error vector (δI_(k), δQ_(k)) is the distance froman ideal position to an actual position of the received complex valuesand S is the magnitude of the vector to an ideal constellation point.24. The communication device of claim 23, if the PHY layer usesπ/2-DBPSK modulation, the EVM is equal to or less than 20%.
 25. Thecommunication device of claim 23, if the PHY layer uses π/4-DQPSKmodulation, the EVM is equal to or less than 12.5%.
 26. Thecommunication device of claim 23, if the PHY layer uses π/8-D8PSKmodulation, the EVM is equal to or less than 7.0%.
 27. The communicationdevice of claim 23 wherein EVM is measured on baseband I and Q samplesafter a received signal is passed through a reference receiver thatperforms the following operations: matched SRRC filtering,carrier-frequency offset estimation and symbol timing recovery whilemaking the measurements.
 28. The communication device of claim 1wherein, for a packet error rate (PER) of less than 10% with a PSDU of255 bytes, minimum receiver sensitivity for the 402-405 MHz frequencyband is −98 dBm at 75.9 kbps, −95 dBm at 151.8 kbps, −92 dBm at 303.6kbps, and −86 dBm at 455.4 kbps.
 29. The communication device of claim 1wherein, for a packet error rate (PER) of less than 10% with a PSDU of255 bytes, minimum receiver sensitivity for the 420-450 MHz frequencyband is −93 dBm at 75.9 kbps, −90 dBm at 151.8 kbps, and −87 dBm at187.5 kbps.
 30. The communication device of claim 1 wherein, for apacket error rate (PER) of less than 10% with a PSDU of 255 bytes,minimum receiver sensitivity for the 863-870 MHz frequency band or the950-956 frequency band is −97 dBm at 101.2 kbps, −94 dBm at 202.4 kbps,−90 dBm at 404.8 kbps, and −85 dBm at 607.1 kbps.
 31. The communicationdevice of claim 1 wherein, for a packet error rate (PER) of less than10% with a PSDU of 255 bytes, minimum receiver sensitivity for the902-928 MHz frequency band is −96 dBm at 121.4 kbps, −93 dBm at 242.9kbps, −89 dBm at 485.7 kbps, and −84 dBm at 728.6 kbps.
 32. Thecommunication device of claim 1 wherein, for a packet error rate (PER)of less than 10% with a PSDU of 255 bytes, minimum receiver sensitivityfor the 2360-2400 MHz frequency band or the 2400-2483.5 MHz frequencyband is −95 dBm at 121.4 kbps, −93 dBm at 242.9 kbps, −90 dBm at 485.7kbps, and −86 dBm at 971.4 kbps.
 33. The communication device of claim 1wherein the PHY layer performs clear channel assessment (CCA) accordingto an energy above threshold mode.
 34. The communication device of claim1 wherein the PHY layer performs clear channel assessment (CCA)according to a carrier sense only mode.
 35. The communication device ofclaim 1 wherein the PHY layer performs clear channel assessment (CCA)according to a carrier sense with energy above threshold mode.